A so called emitter-switching circuit configuration comprises a connection of a bipolar transistor having a high breakdown voltage and a power MOSFET transistor with low voltage.
Such a configuration is schematically shown in FIG. 1 and globally indicated at 1. The emitter-switching configuration 1 comprises a bipolar transistor T1 and a MOS transistor M1 inserted, in series to each other, between a first and a second voltage reference, in particular the supply voltage Vcc and the ground GND.
The emitter-switching configuration 1 provides that the bipolar transistor T1 is of the HV (High Voltage) type, i.e. a transistor having a high breakdown voltage, while the MOS transistor M1 is of the LV (Low Voltage) type, i.e. a transistor having a low breakdown voltage.
The bipolar transistor T1 has a collector terminal connected to the supply voltage reference Vcc through an inductive load L1 and a control or base terminal connected to a driving circuit 2.
The MOS transistor M1 has in turn a control or gate terminal connected to the driving circuit 2.
The driving circuit 2 comprises:                a first resistive element RB connected to the base terminal of the bipolar transistor T1 and, through a Zener diode DZ, to the ground GND;        a second resistive element RG connected to the gate terminal of the MOS transistor M1 and, through a generator of voltage pulses G1, to the ground GND; and        an electrolytic capacitor CB connected, in parallel, to the Zener diode DZ and having, at its ends, a voltage value equal to VB.        
In particular, the electrolytic capacitor CB has the task of storing energy during the turn-off of the bipolar transistor T1, to use it again during a successive turn-on and conduction step of the transistor itself, while the Zener diode prevents the value of the base voltage of the bipolar transistor T1 from exceeding a determined threshold.
The set of the emitter-switching configuration 1 and of the driving circuit 2 forms a device of the controlled emitter-switching type 5.
The emitter-switching configuration, known to the skilled person in the field for a long time, is currently particularly interesting due to the presence on the market of bipolar transistors having a squared RBSOA [Reverse Biased Safe Operating Area] (in emitter-switching configuration) at a current next to the peak one and at a voltage equal to the breakdown voltage BVCES [Breakdown Voltage Collector-Emitter Short], the voltage between the collector and emitter terminals when the base terminal is short-circuited with the emitter terminal, as well as of power MOS transistors having a very low drain-source resistance value when conductive [ON condition], RDSON, and therefore almost assimilable to ideal switches.
Advantages of the emitter-switching configuration are a very low voltage drop in conduction (typical of bipolar transistors) and a high turn-off speed.
During the turn-off, in fact, the current outgoing from the base terminal of the bipolar transistor of the emitter-switching configuration is equal to the current of the collector terminal of this transistor, i.e. a current having a very high value. This determines a drastic reduction both of the storage time and of the fall time, allowing the emitter-switching configuration to work also up to frequencies of 150 kHz.
The driving carried out by means of the driving circuit 2 is very useful and efficient in all those cases wherein the current in the emitter-switching configuration 1 is null, or very small with respect to the nominal one, in the turn-on phase.
In order for this driving to be efficient it is, however, often necessary that the value of the base current during the turn-off phase, IBOFF, multiplied by the storage time, tstorage, is next to the value of the base current in the conduction phase, IBON, multiplied by the turn-on time, ton, i.e.:IBOFF*tstorage>>IBON*tON  (1)
The condition (1) usually occurs when working at relatively high frequencies and with not too high currents, or better when the gain value Hfe of the bipolar transistor is not too low.
In fact, in this case, the driving energy necessary for the conduction is only a little higher with respect to the one recovered during the turn-off phase. It is thus enough to supply the base terminal with a very little power for replacing the unavoidable losses.
FIG. 2 shows the pattern of the values of the voltage VGS between gate and source terminals of the MOS transistor M1, of the voltage between the collector terminal of the bipolar transistor T1 and the source terminal of the MOS transistor M1, VCS, and of the base and collector currents of the bipolar transistor T1 with reference to a flyback converter working at a frequency of 100 kHz and having a null turn-on current since the converter works in a discontinuous way.
When working with applications wherein the value of the current on the device at the turn-on is not null, and at relatively high frequencies (>60 kHz), having to deal with a bipolar device, the phenomenon of the dynamic VCESAT is highlighted. This phenomenon is that, at the turn-on, there exists a certain delay before reaching the value of static voltage VCESAT, it is thus necessary to flood by carriers the base region of the bipolar transistor as quickly as possible to make the value of the voltage VCESAT decrease and to reach the steady value as soon as possible.
For this reason, with the driving circuit 2 of FIG. 1, an excessive dissipation at the turn-ON would be obtained due to the fact that the voltage drop VCE would take a relatively long time (≧2 μs) to reach the saturation value.
Always with reference to the controlled emitter-switching device 5 of FIG. 1, an increase of the VB would only partially reduce the problem of the dynamic VCESAT but it would enormously worsen the performances at the turn-off.
The driving circuit suitable for the applications with non-null collector current at the turn-on has been object of the European patent application No. 03425140.5 (which is incorporated by reference) and it is schematically shown in the annexed FIG. 3, globally indicated at 12. The driving circuit 12 is suitably connected to an emitter-switching configuration 1 to form a device of the controlled emitter-switching type 15.
The driving circuit 12 suitably modulates the base current optimizing both the switch steps and allowing the attainment of the minimum value of VCESAT in the shortest time possible. In the annexed FIG. 4, the waveforms are reported which refer to a converter of the forward type working at a frequency of 110 kHz, where the modulation of the base current can be observed.
The detailed operation and the sizing of the driving circuit 12 are widely discussed in the cited patent application.
Although advantageous under several points of view, this known driving circuit has admitted drawbacks mainly highlighted when the need of applications with collector currents of variable value in a wide range is to be faced.
In these applications, the driving circuit 12, as described in the above indicated European patent application, is in fact sized taking into account the most stressing working condition, i.e. the highest collector current.
Thus a correct saturation level should be ensured for the highest current value, by suitably choosing a base voltage value Vb′ of the bipolar transistor T1, but in so doing, for low collector current values, the device works in over-saturation conditions, obtaining extremely long storage times.
This implies an excessive dissipation at the turn-off, as well as inaccurate control due to the fact that the effective turn-off of the device occurs with delay with respect to the signal supplied by a controller.
It is also possible to use more complex solutions comprising additional circuitries for the control of the storage time, tstorage, through a modulation of the base current.
In this case, however, specific designs of the base driving circuit are to be provided. The circuits thus obtained become difficult to be used with high currents values and they may not allow the connection of a recirculation diode between the collector and source terminals in half-bridge and complete bridge configurations.
In fact, the inductive recirculation current of interest in this case flows in a base loop through a supply capacity, discharging it and flowing then in the base-collector junction instead of flowing in the recirculation diode having the anode connected to the source terminal and the cathode to the collector terminal.